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application note AN255/1288 a designer's guide to the l200 voltage regulator delivering 2 a at a voltage variable from 2.85 v to 36 v, the l200 voltage regulator is a versatile device that simplifies the design of linear supplies. this design guide describes the operation of the device and its ap- plications. the introduction of integrated regulator circuits has greatly simplified the work involved in designing supplies. regulation and protection circuits required for the supply, previously realized using discrete components, are now integrated in a single chip. this had led to significant cost and space saving as well as increased reliability. today the designer has a wide range of fixed and adjustable, positive and negativeseries regulators to choose from as well as an increasing number of switching regulators. the l200 is a positive variable voltage regulator which includes a current limiter and supplies up to 2 a at 2.85 to 36 v. the output voltage is fixed with two resistors or, if a continuously variable output voltage is required, with one fixed and one variable resistor. the maximum output current is fixed with a low value resistor. the device has all the characteristics common to normal fixed regulators and these are described in the datasheet. the l200 is particularly suitable for applications requiring output voltage variation or when a voltage not provided by the stan- dard regulators is required or when a special limit must be placed on the output current. the l200 is available in two packages : pentawatt - offers easy assembly and good reliabil- ity. the guaranteed thermal resistance (r th j-case )is 3 c/w (typically 2 c/w) while if the device is used without heatsink we can consider a guaranteed junction-ambient thermal resistance of 50 c/w. to-3 - for professional and military use or where good hermeticity is required. the guaranteed junction-case thermal resistance is 4 c/w, while the junction-ambient thermal resis- tance is 35 c/w. the junction-case thermal resistance of this pack- age, which is greater than that of the pentawatt, is partlycompensatedby the lower contactresis-tance with the heatsink, especially when an electrical in- sulator is used. circuit operation as can be seen from the block diagram (fig. 1) the voltage regulation loop is almost identical to that of fixed regulators.the only differenceis thatthe nega- tive feedback network is external, so it can be varied (fig. 3). the output is linked to the reference by : v out =v ref (1+ r2 ) (1) r1 considering v out as the outputof an operationalam- plifier with gain equal to g v = 1 + r2/r1 and input signal equal to v ref , variability of the output voltage can be obtained by varying r1 or r2 (or both). it's best to vary r1 because in this way the current in resistors r1 and r2 remains constant (this current is in fact given by v ref /r1). equation(1) can also be found in anotherway which is more useful in order to understand the descrip- tions of the applications discussed. v out =r1 i 1 +r2i 2 and since in practice i 1 ?i 4 (i 4 has a typical value of 10 m a) we can say that v out +r1 i 1 +r2i 1 with i 1 = v ref r1 therefore v out = r2 v ref +v ref =v ref (1+ r2 ) r1 r1 in other words r1 fixes the value of the current cir- culating in r2 so r2 is determined. 1/21
figure 1 : block diagram. figure 2 : schematic diagram. application note 2/21
figure 3. overload protection the device has an overload protection circuit which limits the current available. referring to fig. 2, r24 operatesas acurrent sensor. when at the terminals of r24 there is a voltage drop sufficient to make q20 conduct,q19 begins to draw current from the base of the power transistor (dar- lington formed by q22 and q23) and the output cur- rent is limited. the limit depends on the current which q21 injects into the base of q20. this current dependson the drop-outand the temperaturewhich explains the trend of the curves in fig. 4. figure 4. thermal protection the junction temperature of the device may reach destructive levels during a short circuit at the output or due to an abnormal increase in the ambient tem- perature. to avoid having to use heatsinks which are costly and bulky, a thermal protection circuit has been introduced to limit the outputcurrent so that the dissipated power does not bring the junction tem- perature above the values allowed. the operation of this circuit can be summarized as follows. in q17 there is a constant current equal to : v ref v be17 (v ref = 2.75 v typ) r17 + r16 the base of q18 is therefore biased at : v be18 = v ref v be17 ? r16 @ 350 mv r16 + r17 therefore at t j =25 c q18 is off (since 600 mv is needed for it to start conducting). since the v be of a silicon transistor decreases by about 2 mv/ c, q18 starts conducting at the junction temperature : t j = 600 350 + 25 = 150 c 2 current limitation the innovativefeature of this device is the possibility of acting on the current regulationloop, i.e. of limiting the maximum current that can be suppliedto the de- sired value by using a simple resistor (r3 in fig. 2). obviously if r3 = 0 the maximum output current is application note 3/21
also the maximum current that the device can sup- ply because of its internal limitation. the current loop consists of a comparator circuit with fixed threshold whose value is v sc . this com- parator intervenes when i o .r3 =v sc ,hence i o = v sc (v sc is the voltage between pin 5 r3 and 2 with typical value of 0.45 v). special attention has been given to the comparator circuit in order to ensure that the device behaves as a current generator with high output impedance. typical applications programmable current regulator fig. 5 shows the device used as current generator. in this case the error amplifier is disabled by short- circuiting pin 4 to ground. figure 5. the output current i o is fixed by means of r : i o = v 52 r the output voltage can reach a maximum value v i v drop @ v i 2v (v drop depends on i o ). programmable voltage regulator fig. 6 shows the device connected as a voltage regulator and the maximum output current is the maximum current that the device can supply. the output voltage v o is fixed using potentiometer r2. the equation which gives the output voltage is as follows : v o =v ref (1 + r2 ) r1 by substituting the potentiometer with a fixed resis- tor and choosing suitable values for r1 and r2, it is possible to obtain a wide range of fixed output volt- ages. figure 6. the following formulas and tables can be used to calculate some of the most common output volt- ages. having fixed a certain v o , using the previous for- mula, the maximum value is : v omax =v ref max (1 + r2 max ) and the r1 min minimum value is : v o min =v ref min (1 + r2 min ) r1 max the table below indicates resistor values for typical output voltages : v o 4% r1 1% r2 1% 5v 1.5k w 1.2k w 12v 1k w 3.3kw 15v 750 w 3.3kw 18v 330 w 1.8kw 24v 510 w 3.9k w programmable current and voltage regulator the typical configuration used by the device as a voltage regulator with external current limitation is shown in fig. 7. the fixed voltage of 2.77 v at the ter- minals of r1 makes it possible to force a constant current across variable resistor r2. if r2 is varied, the voltage at pin 2 is varied and so is the output volt- age. i o = v ref ? ? ? 1 + r 2 r 1 ? ? ? i o = v 5 - 2 r application note 4/21
the output voltage is given by : v o =v ref ? (1 + r2 ), with v ref = 2.77 v typ r1 and the maximum output current is given by : i o max = v 52 with v 52 = 0.45 v typ. r3 to maintain a sufficient current for good regulation the value of r1 should be kept low. when there is no load, the output current is v ref /r1. suitable val- ues of r1 are between 500 w and 1.5 k w .iftheload is always present the maximum value for r1 is lim- ited by the current value (10 m a) at the input of the error amplifier (pin 4). figure 7. digitally selected regulator with in- hibit the output voltage of the device can be regulated digitally as shown in fig. 8. the output voltage de- pends on the divider formed by r5 and a combina- tion of r1, r2, r3 and p2. the device can be switched off with a transistor. when the inhibit transistor is saturated, pin 2 is brought to ground potential and the output voltage does not exceed 0.45 v. reducing power dissipation with drop- ping resistor if may sometimes be advisable to reduce the power dissipated by the device. a simple and economic method of doing this is to use a resistor connected in series to the input as shown in fig. 9. the input- output differential voltage on the device is thus re- duced. the formula for calculating r is as follows : r= v imin (v o +v drop ) i o where v drop is the minimum differential voltage be- tween the input and the output of the device at cur- rent i o .v in min is the minimum voltage. v o is the output voltage and i o the output current. with constant load, resistor r can be connected be- tween pins 1 and 2 of the ic instead of in series with the input (fig. 10). in this way, part of the load current flows through the device and part through the resis- tor. this configuration can be used when the mini- mum current by the load is : i omin = v drop (instant by instant) r figure 8. figure 9. i o = v ref ? ? ? 1 + r 2 r 1 ? ? ? i o ( max. ) = v 5 - 2 r application note 5/21
figure 10. soft start when a slow rise time of the output voltage is re- quired, the configuration in fig. 11 can be used. the rise time can be found using the following formula : t on = cv o r 0.45 at switch on capacitor c is discharged and it keeps the voltage at pin 2 low ; or rather, since a voltage of more than 0.45 v cannot be generated between pins 5 and 2, the v o follows the voltage at pin 2 at less than 0.45 v. figure 11. capacitor c is charged by the constant current i c . i c = v sc r therefore the output reaches its nominal value after the time t on : v o v sc = i c ? t on c t on =c ? v o 0.45 ? r @ cv o r 0.45 0.45 light controller fig. 12 shows a circuit in which the output voltage is controlled by the brightnessof the surroundingen- vironment. regulation is by means of a photo resis- tor in parallel with r1. in this case, the output vol-tage increases as the brightness increases. the oppositeeffect, i.e. dimming the light as the ambient light increases, can be obtained by connecting the photoresistor in parallel with r2. figure 12. light dimmer for car display although digital displays in cars are often more aes- thetically pleasing and frequently more easily read they do have a problem. under varying ambient light conditions they are either lost in the background or alternatively appear so bright as to distract the driver. with the system proposed here, this problem is overcome by automatically adjusting the display brightness during daylight conditions and by giving the driver control over the brightness during dusk and darkness conditions. the circuit is shown in fig. 13. the primary supply is shown taken straight from the car battery however it is worth noting that in a car there is always the risk of dump voltages up to 120 v and it is recommended that some form of protectionis included against this. under daylight conditions i.e. with sidelights off and t1 not conducting the output of the device is deter- mined by the values of r1, r2 and the photoresistor (ptr). the output voltage is given by v out =v ref (1 + r2 ) ptr//r1 if the ambient light intensity is high, the resistance of the photoresistorwill be low and therefore v out will be high. as the light decreases, so v out decreases dimming the display to a suitable level. application note 6/21
figure 13. in dusk conditions,when the sidelights are switched on, t1 starts to conductwith its conduction set by the potentiometer wiper at its uppermost position the sidelights are at their brightest and current through t1 would be a minimum. with the wiper at its lowest position obviously the opposite conditions apply. the current through t1 is felt at the summing node a along with the currents through r2 and the parallel network r1, ptr. since v ref is constant the current flowing through r1, ptr must also be constant. therefore any change in the current through t1 causes an equal and opposite change in the current through r2. therefore as i t1 increases, v out de- creases i.e. as the brightness of the side-lights is in- creased or decreased so is the brightness of the display. the values of r2 and ptr should be selected to give the desired minimum and maximum brightness levels desired under both automatic and manual conditions although the minimum brightness under manual conditions can also be set by the maximum current flowing through t1 and, in any case, this should not exceed the maximum current through r2 under automatic operation. the circuit shown with a small modification can also be used for dimmers other than in a car. fig. 15 shows the modification needed. the zener diode should have a v f 2.5 v at i = 10 m a. higher input or output voltages certain applications may require higher input or out- put voltages than the device can produce. the prob- lem can be solved by bringingthe regulator back into figure 14. figure 15. application note 7/21
the normal operating units with the help of external components. when there are high input voltages, the excess vol- tage must be absorbed with a transistor. figs. 16 and 17 show the two circuits : figure 16. figure 17. the designer must take into account the dissipated power and the soa of the preregulation transistor. for example, using the bdx53, the maximum input voltage can reach 56 v (fig. 16). in these conditions we have 20 v of v ce on the transistor and with a load current of 2 a the operation point remains inside the soa. the preregulation used in fig. 16 reduces the ripple at the input of the device, making it possible to obtain an output voltage with negligible ripple. if high output voltages are also required, a second zener, v z , is used to refer the ground pin of an ic to a potential other than zero ; diode d1 provides out- put shortcircuit protection (fig. 18). figure 18. positive and negative voltage regula- tors the circuit in fig. 19 provides positive and negative balanced, stabilized voltages simultaneously. the l200 regulator supplies the positive voltage while the negative is obtained using an operationalampli- fier connected as follower with output current booster. tracking of the positive voltage is achieved by put- ting the non-inverting input to ground and using the inverting input to measure the feedback voltage coming from divider r1-r2. the system is balanced when the inputs of the op- erationalamplifier are at the same voltage, or, since one input is at fixed ground potential, when the vol- tage of the intermediate point of the divider foes to 0 volts. this is only possible if the negative voltage, on command of the op-amp, goes to a value which will make a current equal to that in r1 flows in r2. the ratio which expresses the negative output vol- tage is : v =v + ? r2 (if r2 = r1, we'll get v =v + ) r1 application note 8/21
figure 19. since the maximum supply voltage of the op amp used is 22 v, when pin 7 is connected to point b output voltages up to about 18 v can be obtained. if on the other hand pin 7 is connected to point a, much higher output voltages, up to about 30 v, be obtained since in this case the input voltage can rise to 34 v. fig. 20 shows a diagram is which the l165 power op amp is used to produce the negative voltage. in this case (as in fig. 19) the output voltage is limited by the absolute maximum rating of the supply vol- tage of the l165 which is 18 v. therefore to get a higher vout we must use a zener to keep the device supply within the safety limits. if we have a transformer with two separate secon- daries, the diagram of fig. 21 can be used to obtain independent positive and negative voltages. the two output diodes, d1 and d2, protect the devices from shortcircuits between the positive and negative outputs. figure 20. a : for 18 v v i 32 v note : v z must be chosen in order to verify 2 v i v z =36v b : for v i 18 v a:v i(max) 34 v 3 < v o <30 b:v i(max) 22 v 3 < v o <18 application note 9/21
figure 21. compensation of voltage drop along the wires the diagram in fig. 22 is particularly suitable when a load situated far from the output of the regulator has to be supplied and when we want to avoid the use of two sensing wires. in fact, it is possible to compensate the voltage drop on the line caused by the load current (see the two curves in fig. 23 and 24). r k transforms the load current i l into a propor- tional voltage in series to the reference of the l200. r k i l is then amplified by the factor r2 + r1 r1 with the values of r z , r2 and r1 known, we get : r k =r z r1 r1 + r2 r z , r1 and r2 are assumed to be constant. if r k is higher than 10 w , the output voltage should be calculated as follows : v o =i d r k +v ref r2 + r1 r1 figure 22. application note 10/21
figure 23. figure 24. motor speed control fig. 25 shows how to use the device for the speed control of permanent magnet motors. the desired speed, proportional to the voltage at the terminal of the motor, is obtained by means of r1 and r2. v m =v ref (1 + r2 ) r1 to obtain better compensation of the internal motor resistance, which is essential for good regulation, the following equation is used : r3 r1 ? rm r2 this equation works with infinite r4. if r4 is finite, the motor speed can be increased without altering the ratio r2/r1 and r3. since r4 has a constant voltage (v ref ) at its terminals, which does not vary as r4 varies, this voltage acts on r2 as a constant cur- rent sourcevariable with r4. thevoltage drop on r2 thus increases, and the increase is felt by the volt- age at the terminals of the motor. the voltage in- crease at the motor terminals is : v m = v ref ? r2 r4 + r3 a circuit for a 30 w motor with r m =4 w ,r1=1k w , r2 = 4.3 k w , r4 = 22 k w and r3 = 0.82 w has been realized. power amplitude modulator in the configuration of fig. 26 the l200 is used to send a signal onto a supply line. since the input sig- nal v i is dc decoupled, the v o is defined by : v o =v ref (1 + r2 ) r1 figure 25. the amplified signal v i whose value is : g v = r2 r3 is added to this component. by ignoring the current entering pin 4, we must impose i 1 =i 2 +i 3 (1) and since the voltage between pin 4 and ground remains fixed (v ref ) as long as the device is not in saturation, i 1 = 0 and equation (1) becomes : i 2 =i 3 with i 3 = v i (for x c ? r3) therefore r3 v o =r2 i 2 = v i ? r2 r3 an application is shown in fig. 27. if the dc level is to be varied but not the ac gain, r1 should be re- placed by a potentiometer. application note 11/21
figure 26. figure 27. high current regulators to get a higher current than can be supplied by a sin- gle device one or more external power transistors must be introduced. the problem is then to extend all the device's protection circuits (short-circuit pro- tection, limitation of t j of externalpower devices and overload protection)to the externaltransistors. con- stant current or foldback current limitation therefore becomes necessary. when the regulator is expected to withstand a per- manent shortcircuit, constant current limitation be- comes more and more difficult to guarantee as the nominal v o increases. this is because of the in- crease in v ce at the terminals of the transistor, which leads to an increase in the dissipated power. the heatsinkhas to becalculated in the heaviestworking conditions, and therefore in shortcircuit. this in- creases weight, volume and cost of the heatsink and increase of the ambient temperature (because of high power dissipation). besides heatsink, power transistors must be dimensioned for the short-cir- cuit. this type, of limitation is suited, for example, with highly capacitiveloads. efficiency is increased if pre- regulation is used on the input voltage to maintain a constant drop-out on the power element for all v out , even in shortcircuit. foldback limitation, on the other hand, allows lighter shortcircuit operatingconditions than the previous case. the type of load is impor- tant. if the load is highly capacitive, it is not possible to have a high ratio between i max and i sc because at switch-on, with load inserted, the output may not reach its nominal value. other protection against input shortcircuit, mains failure, overvoltagesand output reverse bias can be realized using two diodes, d1 and d2, inserted as indicated in fig. 28. application note 12/21
figure 28. use of a pnp transistor fig. 29 shows the diagram of a high current supply using the current limitation of the l200. the output current is calculated using the following formula : i o = vsc @ 0.45 v = 4.5 a rsc 0.1 w constant current limitation is used ; so, in output shortcircuit conditions, the transistor dissipates a power equal to : p d =v i ? i o =v i ? v sc r sc the operating point of the transistor should be kept well within the soa ; with r sc = 0.1 w ,v i must not exceed 20 v. part of the i o crosses the transistor and part crosses the regulator. figure 29. the latter is given by : i reg =i b + v be r where i b is the base current of the transistor ( 100 ma at i c =4a)andv be is the base-emitter volt- age ( 1 v at i c = 4 a) ; with r = 2.5 w ,i reg @ 500 ma. use of an npn power transistor fig. 30 shows the same application as described in figure 29, using an npn power transistor instead of a pnp. in this case an externalsignal transistor must be used to limit the current. therefore : i o = v be q1 r sc as regards the output shortcircuit, see par. 1.5. figure 30. application note 13/21
12v - 4a power supply the diagram in fig. 31 shows a supply using the l200 and the bd705. the 1 k w potentiometer,pt1, togetherwith the 3.3 k resistor areused for fine regu- lation of the output voltage. current limitation is of the type shown in fig. 32. trimmer pt2 acts on strech ab of characteristic. with the values indicated (pt2 = 1 k w ,pt3=470 w , r=3k w ), currents from 3 to 4 a can be limited. the field of variation can be increased by increasing the value of r sc or by connecting one terminal of pt to the base of the power transistor, which, however, provides less stable limitation. if section ab is moved, section bc will also be moved. the slope of bc can be varied using pt3. the vol- tage level at point b is fixed by the voltage of the zener diode. the capacitor in parallel to the zener ensures correct switch-on with full load. the bd705 should always be used well within its safe operating area. if this is not possible two or more bd705s should be used, connected in parallel (fig. 33). further protection for the external power transistor can be provided as shown in fig. 34. the ptc resis- tor, whose temperature interventionpoint must pre- vent the t j of the power transistor from reaching its maximum value, should be fixed to the dissipator near the power transistor. dimensioning of r a and r b depends on the ptc used. figure 31. application note 14/21
figure 32. figure 33. figure 34. voltage regulator from 0v to 16v - 4.5a fig. 35 shows an application for a high current sup- ply with output voltage adjustable from 0 v to 16 v, realized with two l200 regulators and an external power transistor. with the values indicated, the cur- rent can be regulated from 2 a to 4.5 a by potenti- ometer pt2. pt1, on the other hand,is used for con- stant current or foldback current limitation. the inte- grated circuit ic2, which does not require a heatsink and has excellent temperature stability, is used to obtain the 0 v output. it is connected so as to lower pin 3 of ic1 until pin 4 reaches 0 v. q1 and q2 en- sure correct operationof the supply at switch-on and switch-off. application note 15/21
figure 35. power supply with v o = 2.8 to 18 v, i o =0to 2.5 a the diagram in fig. 36 shows a supply with output voltage variable from 2.8 v to 18 v and constant cur- rent limitation from 0 a to 2.5 a. the output current can be regulated over a wide range by means of the op. amp. and signal transistor tr 2 . the op. amp. and the transistor are connected in the voltage-cur- rent converter configuration. the voltage is taken at the terminals of r3 and converted into current by pt 2 . i o is fixed as follows : r4 i o =i 1 (*) (**) i sc = v sc pt2 r2 when i 1 =i sc , the regulatorstarts to operate as a cur- rent generator. by making (*) equal to (**) we get : r4 i o = v sc ; therefore i o = vsc ? pt 2 pt 2 r2 r2 ? r4 diodes d1 and d2 keep transistor tr 2 in linear con- dition in the case of small output currents. if it is not necessary to limit the current to zero, one of the di- odes can be eliminated : the second diode could also be eliminated if tr 1 were a darlington instead of a transistor. the op. amp. must have inputs compatible with ground in order to guarantee current limitation even in shortcircuit. with a negative voltage available, even of only &a few volts, current limitation is sim- plified. application note 16/21
figure 36. layout considerations the performance of a regulator depends to a great extent on the case with which the printed circuit is produced. there must be no impulsive currents (like the one in the electrolytic filter capacitor at the input of the regulator) between the ground pin of the de- vice (pin 3) and the negative output terminal be- cause these would increase the output ripple. care must also be taken when inserting the resistor con- nected between pin 4 and pin 3 of the device. the track connecting pin 3 to a terminal of this re- sistor should be very short and must not be crossed by the load current (which, since it is generally vari- able, would give rise to a voltage drop on this stretch of track, altering the value of v ref and thereforeof v o . when the load is not in the immediate proximity of the regulator output o+ senseo and o senseo termi- nals should be used (see fig. 37). by connectingthe o+ senseo and o senseo terminals directly at the charge terminals the voltage drop on the connection cable between supply and load are compensated. fig. 37 shows how to connect supply and load using the sensing clamps terminals. figure 37. application note 17/21
figure 38. heatsink dimensioning the heatsink dissipates the heat produced by the device to prevent the internal temperature from reaching value which could be dangerous for device operation and reliability. integrated circuits in plastic packagemust never ex- ceed 150 c even in the worst conditions. this limit has been set because the encapsulating resin has problems of vitrification if subjected to temperatures of more than 150 c for long periods or of more than 170 c for short periods (24 h). in any case the tem- perature accelerates the ageing process and there- fore influences the device life ; an increase of 10 c can halve the device life. a well designed heatsink should keep the junction temperature between 90 c and 110 c. fig. 39 shows the structure of a power device. as demonstratedin thermodynamics, a thermal circuit can be considered to be an electri- cal circuit where r1, 2 represent the thermal resis- tance of the single elements (expressed in c/w) ; figure 40. figure 39. c1, 2 the thermal capacitance (expressed in c/w) i the dissipated power v the temperature difference with respect to the reference (ground) this circuit can be simplified as follows : figure 41. where c e is the thermal capacitanceof the die plus that of the tab. c h is the thermal capacitance of the heatsink r jc is the junction case thermal resistance r h is the heatsink thermal resistance but since the aim of this section is not that of studing the transistors, the circuit can be further reduced. application note 18/21
figure 42. if we now consider the ground potential as ambient temperature, we have : t i =t a +(r jc +r h )p d (1) rth = t i t a r ic ? p d (1a) pd t c =t a +r h ? p d (2) for example, consider an application of the l200 with the following characteristics : v in typ =20v v o =14v typical conditions io typ = 1 a t a =40 c v in max =22 v v o =14v overload conditions i omax = 1.2 a t a =60 c p dtyp =(v in v o ) ? i o = (2014) ? 1=6w p d max = (2214) ? 1.2 = 9.6 w imposing t j =90 c of (1a) we get (from l200 char- acteristics we get r jc =3 c/w). r h = 90 40 3 ? 6 =5.3 c/w 6 using the value thus obtained in (1), we get that the junctiontemperatureduring the overload goes to the following value : t j = 60 + (3 + 5.3) . 9.6 = 140 c if the overload occurs only rarely and for short peri- ods, dimensioning can be considered to be correct. obviously during the shortcircuit, the dissipated power reaches must higher values (about 40 w for the case considered) but in this case the thermal protection intervenes to maintain the temperature below the maximum values allowed. note 1 : if insulating materials are used between de- vice and heatsink, the thermal contact resistance must be taken into account (0.5 to 1 c/w, depend- ing on the type of insulant used) and the circuit in fig. 43 becomes : figure 43. note 2 : in applications where one or more external transistors are used together with the l200, the dis- sipated power must be calculated for each compo- nent. the various junction temperatures can be calculated by solving the following circuit : figure 44. this applies if the various dissipating elements are fairly near to one another with respect to the heatsink dimensions, otherwise the heatsink can no longer be considered as a concentrated constant and the calculation becomes difficult. this concept is better explained by the graph in fig. 45 which shows the case (and therefore junction) temperature variation as a function of the distance between two dissipating elements with the same type of dissipator and the same dissipated power. the graph in fig. 45 refers to the specific case of two elements dissipating the same power, fixed on a rectangular aluminium plate with a ratio of 3 bet- ween the two sides. the temperature jump will de- pend on the dissipated power and one the device geometry but we want to show that there exists an optimal position between the two devices : d= 1 ? side of the plate 2 fig. 46 shows the trend of the temperatureas a func- tion of the distance between two dissipating ele- application note 19/21
ments whose dissipated power is fairly different (ra- tio 1 to 4). this graph may be useful in applications with the l200 + external transistor (in which the transistor generally dissipates more than the l200) where the temperature of the l200 has to be kept as low as possible and especially where the thermal protec- tion of the l200 is to be used to limit the transistor temperature in the case of an overload or abnormal increase in the ambient temperature. in other words the distance between the two elements can be se- lected so that the power transistor reaches the t j max (200 c for a to-3 transistor) when the l200 reaches the thermal protection intervention tem- perature. figure 45. figure 46. a : positi on of the device with high power dissipation (10 w) b : positi on of the device with low power dissipation (2.5 w) application note 20/21
information furnished is believed to be accurate and reliable. however, sgs-thomson microelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of sgs-thomson microelectronics. specifications mentioned in this publication are subject to change without notice. this publication supersedes and replaces all information previously supplied. sgs-thomson microelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of sgs-thomson microelectronics. ? 1995 sgs-thomson microelectronics - all rights reserved sgs-thomson microelectronics group of companies australia - brazil - france - germany - hong kong - italy - japan - korea - malaysia - malta - morocco - the netherlands - singapore - spain - sweden - switzerland - taiwan - thaliand - united kingdom - u.s.a. application note 21/21

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